Steering diversity for an OFDM-based multi-antenna communication system

ABSTRACT

A transmitting entity uses different steering vectors for different subbands to achieve steering diversity. Each steering vector defines or forms a beam for an associated subband. Any steering vector may be used for steering diversity. The steering vectors may be defined such that the beams vary in a continuous instead of abrupt manner across the subbands. This may be achieved by applying continuously changing phase shifts across the subbands for each transmit antenna. As an example, the phase shifts may change in a linear manner across the subbands for each transmit antenna, and each antenna may be associated with a different phase slope. The application of linearly changing phase shifts to modulation symbols in the frequency domain may be achieved by either delaying or circularly shifting the corresponding time-domain samples.

CLAIM OF PRIORITY UNDER 35 U.S.C. §119

The present Application for Patent claims priority to ProvisionalApplication Ser. No. 60/569,103, entitled “Steering Diversity for anOFDM-Based Multi-Antenna Communication System,” filed May 7, 2004, andassigned to the assignee hereof and hereby expressly incorporated byreference herein.

BACKGROUND

I. Field

The present invention relates generally to communication, and morespecifically to data transmission in a multi-antenna communicationsystem that utilizes orthogonal frequency division multiplexing (OFDM).

II. Background

OFDM is a multi-carrier modulation technique that effectively partitionsthe overall system bandwidth into multiple (K) orthogonal subbands,which are also referred to as tones, subcarriers, bins, and frequencychannels. With OFDM, each subband is associated with a respectivesubcarrier that may be modulated with data. OFDM is widely used invarious wireless communication systems, such as those that implement thewell-known IEEE 802.1a and 802.11g standards. IEEE 802.1a and 802.1ggenerally cover single-input single-output (SISO) operation whereby atransmitting device employs a single antenna for data transmission and areceiving device normally employs a single antenna for data reception.

A multi-antenna communication system may support communication for bothsingle-antenna devices and multi-antenna devices. In this system, amulti-antenna device may utilize its multiple antennas for datatransmission to a single-antenna device. The multi-antenna device andthe single-antenna device may implement any one of a number ofconventional transmit diversity schemes in order to obtain transmitdiversity and improve performance for the data transmission. One suchtransmit diversity scheme is described by S. M. Alamouti in a paperentitled “A Simple Transmit Diversity Technique for WirelessCommunications,” IEEE Journal on Selected Areas in Communications, Vol.16, No. 8, October 1998, pp. 1451-1458. For the Alamouti scheme, thetransmitting device transmits each pair of modulation symbols from twoantennas in two symbol periods, and the receiving device combines tworeceived symbols obtained in the two symbol periods to recover the pairof modulation symbols sent by the transmitting device. The Alamoutischeme as well as most other conventional transmit diversity schemesrequire the receiving device to perform special processing, which may bedifferent from scheme to scheme, in order to recover the transmitteddata and obtain the benefits of transmit diversity.

A “legacy” single-antenna device may be designed for SISO operationonly, as described below. This is normally the case if the wirelessdevice is designed for the IEEE 802.11a or 802.11g standard. Such alegacy single-antenna device would not be able to perform the specialprocessing required by most conventional transmit diversity schemes.Nevertheless, it is still highly desirable for a multi-antenna device totransmit data to the legacy single-antenna device in a manner such thatgreater reliability and/or improved performance can be achieved.

There is therefore a need in the art for techniques to achieve transmitdiversity in an OFDM-based system, especially for legacy single-antennadevices.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a multi-antenna system with an access point and userterminals.

FIG. 2 shows a block diagram of a multi-antenna transmitting entity, asingle-antenna receiving entity, and a multi-antenna receiving entity.

FIG. 3 shows an OFDM waveform in the frequency domain.

FIG. 4 shows a block diagram of an OFDM modulator.

FIG. 5 shows a model for transmission with steering diversity for onesubband.

FIG. 6 shows a transmit (TX) spatial processor and an OFDM modulator.

FIG. 7 shows plots of linear phase shifts across subbands for fourantennas.

FIGS. 8A and 8B show two embodiments for achieving linear phase shiftsusing different delays for time-domain samples.

FIG. 8C shows transmissions from T transmit antennas for the embodimentsshown in FIGS. 8A and 8B.

FIG. 9A shows an embodiment for achieving linear phase shifts usingcircular shifts for time-domain samples.

FIG. 9B shows transmissions from T transmit antennas for the embodimentshown in FIG. 9A.

DETAILED DESCRIPTION

The word “exemplary” is used herein to mean “serving as an example,instance, or illustration.” Any embodiment described herein as“exemplary” is not necessarily to be construed as preferred oradvantageous over other embodiments.

FIG. 1 shows a multi-antenna system 100 with an access point (AP) 110and user terminals (UTs) 120. An access point is generally a fixedstation that communicates with the user terminals and may also bereferred to as a base station or some other terminology. A user terminalmay be fixed or mobile and may also be referred to as a mobile station,a wireless device, a user equipment (UE), or some other terminology. Fora centralized architecture, a system controller 130 couples to theaccess points and provides coordination and control for these accesspoints.

Access point 110 is equipped with multiple antennas for datatransmission and reception. Each user terminal 120 may be equipped witha single antenna or multiple antennas for data transmission andreception. A user terminal may communicate with the access point, inwhich case the roles of access point and user terminal are established.A user terminal may also communicate peer-to-peer with another userterminal. In the following description, a transmitting entity isequipped with multiple (T) transmit antennas, and a receiving entity maybe equipped with a single antenna or multiple (R) antennas. Amultiple-input single-output (MISO) transmission exists when thereceiving entity is equipped with a single antenna, and a multiple-inputmultiple-output (MIMO) transmission exists when the receiving entity isequipped with multiple antennas.

FIG. 2 shows a block diagram of a multi-antenna transmitting entity 210,a single-antenna receiving entity 250 x, and a multi-antenna receivingentity 250 y in system 100. Transmitting entity 210 may be an accesspoint or a multi-antenna user terminal. Each receiving entity 250 mayalso be an access point or a user terminal.

At transmitting entity 210, a transmit (TX) data processor 212 processes(e.g., encodes, interleaves, and symbol maps) traffic/packet data andgenerates data symbols. As used herein, a “data symbol” is a modulationsymbol for data, a “pilot symbol” is a modulation symbol for pilot(which is data that is known a priori by both the transmitting andreceiving entities), a “transmit symbol” is a symbol to be sent from atransmit antenna, and a “received symbol” is a symbol obtained from areceive antenna. A TX spatial processor 220 receives and demultiplexespilot and data symbols onto the proper subbands, performs spatialprocessing as appropriate, and provides T streams of transmit symbolsfor the T transmit antennas. An OFDM modulator (Mod) 230 performs OFDMmodulation on the T transmit symbol streams and provides T streams ofsamples to T transmitter units (TMTR) 232 a through 232 t. Eachtransmitter unit 232 processes (e.g., converts to analog, amplifies,filters, and frequency upconverts) its transmit symbol stream andgenerates a modulated signal. Transmitter units 232 a through 232 tprovide T modulated signals for transmission from T antennas 234 athrough 234 t, respectively.

At single-antenna receiving entity 250 x, an antenna 252 x receives theT transmitted signals and provides a received signal to a receiver unit(RCVR) 254 x. Receiver unit 254 x performs processing that iscomplementary to the processing performed by transmitter units 232 andprovides a stream of samples. An OFDM demodulator (Demod) 260 x performsOFDM demodulation on the sample stream to obtain received data and pilotsymbols, provides the received data symbols to a detector 270 x, andprovides the received pilot symbols to a channel estimator 284 x withina controller 280 x. Channel estimator 284 x derives channel estimatesfor the effective SISO channels between transmitting entity 210 andreceiving entity 250 x for subbands used for data transmission. Detector270 x performs detection on the received data symbols for each subbandbased on the effective SISO channel estimate for that subband andprovides a stream of detected symbols for all subbands. A receive (RX)data processor 272 x then processes (e.g., symbol demaps, deinterleaves,and decodes) the detected symbol stream and provides decoded data.

At multi-antenna receiving entity 250 y, R antennas 252 a through 252 rreceive the T transmitted signals, and each antenna 252 provides areceived signal to a respective receiver unit 254. Each receiver unit254 processes a respective received signal and provides a sample streamto an associated OFDM demodulator 260. Each OFDM demodulator 260performs OFDM demodulation on its sample stream to obtain received dataand pilot symbols, provides the received data symbols to an RX spatialprocessor 270 y, and provides the received pilot symbols to a channelestimator 284 y within a controller 280 y. Channel estimator 284 yderives channel estimates for the actual or effective MIMO channelsbetween transmitting entity 210 and receiving entity 250 y for subbandsused for data transmission. Controller 280 y derives spatial filtermatrices based on the MIMO channel estimates. RX spatial processor 270 yperforms receiver spatial processing (or spatial matched filtering) onthe received data symbols for each subband with the spatial filtermatrix derived for that subband and provides detected symbols for thesubband. An RX data processor 272 y then processes the detected symbolsfor all subbands and provides decoded data.

Controllers 240, 280 x, and 280 y control the operation of theprocessing units at transmitting entity 210 and receiving entities 250 xand 250 y, respectively. Memory units 242, 282 x, and 282 y store dataand/or program code used by controllers 240, 280 x, and 280 y,respectively.

FIG. 3 shows an OFDM waveform in the frequency domain. OFDM provides Ktotal subbands, and the subcarrier for each subband may be individuallymodulated with data. Of the K total subbands, N_(D) subbands may be usedfor data transmission, N_(P) subbands may be used for pilottransmission, and the remaining N_(G) subbands may be unused and serveas guard subbands, where K=N_(D)+N_(P)+N_(G). For example, 802.11autilizes an OFDM structure that has 64 total subbands, of which 48subbands are used for data transmission, 4 subbands are used for pilottransmission, and 12 subbands are unused. In general, system 100 mayutilize any OFDM structure with any number of data, pilot, guard, andtotal subbands. For simplicity, the following description assumes thatall K subbands are usable for data and pilot transmission.

FIG. 4 shows a block diagram of OFDM modulator 230 at transmittingentity 210. The data to be transmitted (or information bits) istypically first encoded to generate code bits, which are theninterleaved. The interleaved bits are then grouped into B-bit binaryvalues, where B≧1. Each B-bit value is then mapped to a specificmodulation symbol based on a modulation scheme selected for use (e.g.,M-PSK or M-QAM, where M=2^(B)). Each modulation symbol is a complexvalue in a signal constellation for the selected modulation scheme. Ineach OFDM symbol period, one modulation symbol may be transmitted oneach subband. (A signal value of zero, which is also called a zerosymbol, is usually provided for each unused subband.) An inversediscrete Fourier transform (IDFT) unit 432 receives K modulation symbolsfor the K subbands in each OFDM symbol period, transforms the Kmodulation symbols to the time domain with a K-point IDFT, and providesa “transformed” symbol that contains K time-domain samples. Each sampleis a complex-value to be transmitted in one sample period. Aparallel-to-serial (P/S) converter 434 serializes the K samples for eachtransformed symbol. A cyclic prefix generator 436 then repeats a portion(or C samples) of each transformed symbol to form an OFDM symbol thatcontains K+C samples. The cyclic prefix is used to combat inter-symbolinterference (ISI) caused by frequency selective fading, which is afrequency response that varies across the overall system bandwidth. AnOFDM symbol period (which is also referred to herein as simply a “symbolperiod”) is the duration of one OFDM symbol and is equal to K+C sampleperiods.

In system 100, a MISO channel exists between a multi-antennatransmitting entity and a single-antenna receiving entity. For anOFDM-based system, the MISO channel formed by the T antennas at thetransmitting entity and the single antenna at the receiving entity maybe characterized by a set of K channel response row vectors, each ofdimension 1×T, which may be expressed as:h (k)=[h ₀(k)h ₁(k) . . . h _(T−1)(k)], for k=0, . . . , K−1,   Eq (1)where k is an index for subband and h_(i)(k), for i=0, . . . , T−1,denotes the coupling or complex gain between transmit antenna i and thesingle receive antenna for subband k. For simplicity, the MISO channelresponse h(k) is shown as a function of only subband k and not time.

If the transmitting entity has an accurate estimate of the MISO channelresponse, then it may perform spatial processing to direct a datatransmission toward the receiving entity. However, if the transmittingentity does not have an accurate estimate of the wireless channel, thenthe T transmissions from the T antennas cannot be intelligently adjustedbased on the wireless channel.

When an accurate channel estimate is not available, the transmittingentity may transmit data from its T antennas to the single-antennareceiving entity using steering diversity to achieve transmit diversity,greater reliability, and/or improved performance. With steeringdiversity, the transmitting entity performs spatial processing such thatthe data transmission observes different effective channels across thesubbands used for data transmission. Consequently, performance is notdictated by a bad channel realization. The spatial processing forsteering diversity is also such that the single-antenna receiving entitycan perform the normal processing for SISO operation (and does not needto do any other special processing for transmit diversity) in order torecover the data transmission and enjoy the benefits of transmitdiversity. For clarity, the following description is generally for oneOFDM symbol, and the index for time is omitted.

FIG. 5 shows a model for transmission with steering diversity for onesubband k from multi-antenna transmitting entity 210 to single-antennareceiving entity 250 x. A modulation symbol s(k) to be sent on subband kis spatially processed with T complex weights (or scalar values) v₀(k)through V_(T−1)(k) to obtain T transmit symbols for subband k, which arethen processed and sent from the T transmit antennas. The T transmitsymbols for subband k observe channel responses of h₀(k) throughh_(T−1)(k).

The transmitting entity performs spatial processing for each subband kfor steering diversity, as follows:x (k)= v (k)·s(k) , for k=0, . . . , K−1,   Eq (2)where s(k) is a modulation symbol to be sent on subband k;v (k)=[v ₀(k) v _(T−1)(k) . . . v _(T−1)(k)]^(T) is a T×1 steeringvector for subband k;x (k)=[x ₀(k) x ₁(k) . . . x _(T−1)(k)]^(T) is a T×1 vector with Ttransmit symbols to be

-   -   sent from the T transmit antennas on subband k; and “^(T)”        denotes a transpose.        In general, the modulation symbol s(k) may be any real or        complex value (e.g., a signal value of zero) and does not need        to be from a signal constellation.

The received symbols at the receiving entity for each subband k may beexpressed as: $\begin{matrix}\begin{matrix}{{{r(k)} = {{{\underset{\_}{h}(k)} \cdot {\underset{\_}{x}(k)}} + {n(k)}}},} \\{{= {{{\underset{\_}{h}(k)} \cdot {\underset{\_}{v}(k)} \cdot {s(k)}} + {n(k)}}},\quad{{{for}\quad k} = 0},\ldots\quad,{K - 1},} \\{{= {{{h_{eff}(k)} \cdot {s(k)}} + {n(k)}}},}\end{matrix} & {{Eq}\quad(3)}\end{matrix}$where r(k) is a received symbol for subband k;

-   -   h_(eff)(k) is an effective SISO channel response for subband k,        which is h_(eff)(k)=h(k)·v(k); and    -   n(k) is the noise for subband k.

As shown in equation (3), the spatial processing by the transmittingentity for steering diversity results in the modulation symbol s(k) foreach subband k observing the effective SISO channel response h_(eff)(k),which includes the actual MISO channel response h(k) and the steeringvector v(k) for that subband. The receiving entity can estimate theeffective SISO channel response h_(eff)(k), for example, based on pilotsymbols received from the transmitting entity. The receiving entity canthen perform detection or matched filtering on the received symbol r(k)for each subband k with the effective SISO channel response estimateĥ_(eff)(k) for that subband to obtain a detected symbol ŝ(k), which isan estimate of the modulation symbol s(k) transmitted on the subband.

The receiving entity may perform matched filtering as follows:$\begin{matrix}{{{\hat{s}(k)} = {\frac{{{\hat{h}}_{eff}^{*}(k)} \cdot {r(k)}}{{{{\hat{h}}_{eff}(k)}}^{2}} = {{s(k)} + {n^{\prime}(k)}}}},} & {{Eq}\quad(4)}\end{matrix}$where “*” denotes a conjugate and n′(k) is the noise after the matchedfiltering. The detection operation in equation (4) is the same as wouldbe performed by the receiving entity for a SISO transmission. However,the effective SISO channel response estimate, ĥ_(eff)(k), is used fordetection instead of a SISO channel response estimate, ĥ(k).

For steering diversity, the receiving entity does not need to knowwhether a single antenna or multiple antennas are used for datatransmission and also does not need to know the steering vector used foreach subband. The receiving entity can nevertheless enjoy the benefitsof transmit diversity if different steering vectors are used across thesubbands and different effective SISO channels are formed for thesesubbands. A data transmission sent across multiple subbands would thenobserve an ensemble of different effective SISO channels across thesubbands used for data transmission.

FIG. 6 shows a block diagram of a TX spatial processor 220 a and an OFDMmodulator 230 a, which are an embodiment of TX spatial processor 220 andOFDM modulator 230, respectively, in FIG. 2. TX spatial processor 220 areceives K modulation symbols (or generically, input symbols) s(0)through s(K−1) for the K subbands for each OFDM symbol period. Within TXspatial processor 220 a, a different set of K multipliers 620 multipliesthe K modulation symbols with a set of K weights v_(i)(0) throughv_(i)(K−1) for each transmit antenna i and provides K weighted symbolsfor that antenna. The modulation symbol s(k) for each subband k istransmitted from all T antennas and is multiplied with T weights v₀(k)through v_(T−1)(k) for the T transmit antennas for that subband. TXspatial processor 220 a provides T sets of K weighted symbols for the Ttransmit antennas.

Within OFDM modulator 230 a, the set of K weighted symbols for eachtransmit antenna i is transformed to the time-domain by a respectiveIDFT unit 632 to obtain a transformed symbol for that antenna. The Ktime-domain samples for the transformed symbol for each transmit antennai are serialized by a respective P/S converter 634 and further appendedwith a cyclic prefix by a cyclic prefix generator 636 to generate anOFDM symbol for that antenna. The OFDM symbol for each transmit antennai is then conditioned by transmitter unit 232 for that antenna andtransmitted via the antenna.

For steering diversity, the transmitting entity uses different steeringvectors for different subbands, with each steering vector defining orforming a beam for the associated subband. In general, it is desirableto use as many different steering vectors as possible across thesubbands to achieve greater transmit diversity. For example, a differentsteering vector may be used for each of the K subbands, and the set of Ksteering vectors used for the K subbands may be denoted as {v(k)}. Foreach subband, the steering vector may be the same over time or maychange, e.g., from symbol period to symbol period.

In general, any steering vector may be used for each of the K subbandsfor steering diversity. However, to ensure that performance is notdegraded for single-antenna devices that are not aware of the steeringdiversity being performed and further rely on some correlation acrossthe subbands, the steering vectors may be defined such that the beamsvary in a continuous instead of abrupt manner across the subbands. Thismay be achieved by applying continuously changing phase shifts acrossthe subbands for each transmit antenna. As an example, the phase shiftsmay change in a linear manner across the subbands for each transmitantenna, and each antenna may be associated with a different phaseslope, as described below. The application of linearly changing phaseshifts to modulation symbols in the frequency domain may be achieved bytemporally modifying (e.g., either delaying or circularly shifting) thecorresponding time-domain samples. If different steering vectors areused for different subbands, then the modulation symbols for thesesubbands are beamed in different directions by the array of N transmitantennas. If encoded data is spread over multiple subbands withdifferent steering, then decoding performance will likely improve due tothe increased diversity.

If the steering vectors for adjacent subbands generate beams in verydifferent directions, then the effective SISO channel responseh_(eff)(k) would also vary widely among the adjacent subbands. Somereceiving entities may not be aware of steering diversity beingperformed, such as legacy single-antenna devices in an IEEE 802.11asystem. These receiving entities may assume that the channel responsevaries slowly across the subbands and may perform channel estimation ina manner to simplify the receiver design. For example, these receivingentities may estimate the channel response for a subset of the K totalsubbands and use interpolation or some other techniques to deriveestimates of the channel response for the other subbands. The use ofabruptly changing steering vectors (e.g., pseudo-random steeringvectors) may severely degrade the performance of these receivingentities.

To provide transmit diversity and avoid degrading the performance oflegacy receiving entities, the steering vectors may be selected suchthat (1) different beams are used for different subbands and (2) thebeams for adjacent subbands have smooth instead of abrupt transitions.The weights to use for the K subbands of the T transmit antennas may beexpressed as: $\begin{matrix}{\underset{\_}{V} = {\begin{bmatrix}{\underset{\_}{v}(0)} & {\underset{\_}{v}(1)} & \ldots & {\underset{\_}{v}\left( {K - 1} \right)}\end{bmatrix} = {\quad{\begin{bmatrix}{v_{0}(0)} & {v_{0}(1)} & \ldots & {v_{0}\left( {K - 1} \right)} \\{v_{1}(0)} & {v_{1}(1)} & \ldots & {v_{1}\left( {K - 1} \right)} \\\vdots & \vdots & ⋰ & \vdots \\{v_{T - 1}(0)} & {v_{T - 1}(1)} & \ldots & {v_{T - 1}\left( {K - 1} \right)}\end{bmatrix},}}}} & {{Eq}\quad(5)}\end{matrix}$where V is a T×K matrix of weights for the K subbands of the T transmitantennas.

In an embodiment, the weights in the matrix V are defined as follows:$\begin{matrix}{{{v_{i}(k)} = {{B(i)} \cdot {\mathbb{e}}^{j\frac{2{\pi \cdot {\mathbb{i}} \cdot k}}{K}}}},{{{for}\quad i} = 0},\ldots\quad,{{T - {1\quad{and}\quad k}} = 0},\ldots\quad,{K - 1},} & {{Eq}\quad(6)}\end{matrix}$where B(i) is a complex gain for transmit antenna i;

-   -   v_(i)(k) is the weight for subband k of transmit antenna i; and    -   j is the imaginary value defined by j={square root}{square root        over (−1)}.

The magnitude of the complex gain for each transmit antenna may be setto one, or ∥B(i)∥=1.0 for i=0, . . . , T−1. The weights shown inequation (6) correspond to a progressive phase shift for each subbandand antenna. These weights effectively form a slightly different beamfor each subband for a linear array of T equally spaced antennas.

In a specific embodiment, the weights are defined as follows:$\begin{matrix}{{{v_{i}(k)} = {{{\mathbb{e}}^{{- {j\pi}} \cdot {\mathbb{i}}} \cdot {\mathbb{e}}^{j\frac{2{\pi \cdot {\mathbb{i}} \cdot k}}{K}}} = {\mathbb{e}}^{{j2\pi}\quad\frac{\mathbb{i}}{K}{({k - \quad\frac{K}{2}})}}}},} & {{Eq}\quad(7)}\end{matrix}$for i=0, . . . , T−1 and k=0, . . . , K−1. The embodiment shown inequation (7) uses B(i)=e^(−jπ×i) for equation (6). This results in adifferent phase shift being applied to each antenna.

FIG. 7 shows plots of the phase shifts for each transmit antenna for acase with T=4. The center of the K subbands is typically considered tobe at zero frequency, as shown in FIG. 3. The weights generated based onequation (7) may be interpreted as creating a linear phase shift acrossthe K subbands. Each transmit antenna i, for i=0, . . . , T−1, isassociated with a phase slope of 2π·i/K. The phase shift for eachsubband k, for k=0, . . . , K−1, for each transmit antenna i is given as2π·i·(k−K/2)/K. The use of B(i)=e^(−jπ×i) result in subband k=K/2observing a phase shift of zero.

The weights derived based on equation (7) may be viewed as a linearfilter having a discrete frequency response of G_(i)(k′), which may beexpressed as: $\begin{matrix}{{{G_{i}\left( k^{\prime} \right)} = {{v_{i}\left( {k^{\prime} + {K/2}} \right)} = {\mathbb{e}}^{{j2\pi}\quad\frac{{\mathbb{i}} \cdot k^{\prime}}{K}}}},} & {{Eq}\quad(8)}\end{matrix}$for i=0, . . . , T−1 and k′=(−K/2), . . . , (K/2−1). The subband index kis for a subband numbering scheme that places the zero frequency atsubband N_(center)=K/2, as shown in FIG. 3. The subband index k′ is ashifted version of the subband index k by K/2, or k′=k−K/2. This resultsin subband zero being at zero frequency for the new subband numberingscheme with the index k′. N_(center) may be equal to some other valueinstead of K/2 if the index k is defined in some other manner (e.g.,k=1, . . . , K) or if K is an odd number.

A discrete time-domain impulse response g_(i)(n) for the linear filtermay be obtained by performing a K-point IDFT on the discrete frequencyresponse G_(i)(k′). The impulse response g_(i)(n) may be expressed as:$\begin{matrix}\begin{matrix}{{{g_{i}(n)} = {\frac{1}{K} \cdot {\sum\limits_{k^{\prime} = {{- K}/2}}^{{K/2} - 1}{{G_{i}\left( k^{\prime} \right)} \cdot {\mathbb{e}}^{{j2\pi}\quad\frac{n \cdot k^{\prime}}{K}}}}}},} \\{{= {\frac{1}{K} \cdot {\sum\limits_{k^{\prime} = {{- K}/2}}^{{K/2} - 1}{{\mathbb{e}}^{{j2\pi}\quad\frac{{\mathbb{i}} \cdot k^{\prime}}{K}} \cdot {\mathbb{e}}^{{j2\pi}\quad\frac{n \cdot k^{\prime}}{K}}}}}},} \\{{= {\frac{1}{K} \cdot {\sum\limits_{k^{\prime} = {{- K}/2}}^{{K/2} - 1}{\mathbb{e}}^{{j2\pi}\quad\frac{k^{\prime}}{K}{({i + n})}}}}},} \\{= \left\{ \begin{matrix}1 & {{{for}\quad n} = {- i}} \\0 & {otherwise}\end{matrix} \right.}\end{matrix} & {{Eq}\quad(9)}\end{matrix}$where n is an index for sample period and has a range of n=0, . . . ,K−1. Equation (9) indicates that the impulse response g_(i)(n) fortransmit antenna i has a single unit-value tap at a delay of i sampleperiods and is zero at all other delays.

The spatial processing with the weights defined as shown in equation (7)may be performed by multiplying the K modulation symbols for eachtransmit antenna i with the K weights v_(i)(0) through v_(i)(K−1) forthat antenna and then performing a K-point IDFT on the K weightedsymbols. Equivalently, the spatial processing with these weights may beachieved by (1) performing a K-point IDFT on the K modulation symbols toobtain K time-domain samples, and (2) performing a circular convolutionof the K time-domain samples with the impulse response g_(i)(n), whichhas a single unit-value tap at a delay of i sample periods.

FIG. 8A shows a block diagram of a TX spatial processor 220 b and anOFDM modulator 230 b, which are another embodiment of TX spatialprocessor 220 and OFDM modulator 230, respectively, in FIG. 2. OFDMmodulator 220 b receives K modulation symbols s(0) through s(K−1) forthe K subbands for each OFDM symbol period. Within OFDM modulator 230 b,an IDFT unit 832 performs a K-point IDFT on the K modulation symbols andprovides K time-domain samples. A P/S converter 834 serializes the Ktime-domain samples. A cyclic prefix generator 836 then appends aC-sample cyclic prefix and provides an OFDM symbol containing K+Csamples to TX spatial processor 220 b. TX spatial processor 220 bincludes T digital delay units 822 a through 822 t for the T transmitantennas. Each delay unit 822 receives and delays the OFDM symbol fromOFDM demodulator 230 b by a different amount determined by theassociated transmit antenna. In particular, delay unit 822 a fortransmit antenna 234 a delays the OFDM symbol by zero sample period,delay unit 822 b for transmit antenna 234 b delays the OFDM symbol byone sample period, and so on, and delay unit 822 t for transmit antenna234 t delays the OFDM symbol by T−1 sample periods. The subsequentprocessing by transmitter units 232 is as described above.

FIG. 8B shows a block diagram of OFDM modulator 230 b and a TX spatialprocessor 220 c, which is yet another embodiment of TX spatial processor220 in FIG. 2. OFDM modulator 220 b performs OFDM modulation on Kmodulation symbols for each OFDM symbol period as described above forFIG. 8A. Transmitter unit 232 then receives and conditions the OFDMsymbol for each symbol period to generate a modulated signal. TX spatialprocessor 220 c provides time delay in the analog domain. TX spatialprocessor 220 c includes T analog delay units 824 a through 824 t forthe T transmit antennas. Each delay unit 824 receives and delays themodulated signal by a different amount determined by the associatedtransmit antenna. In particular, delay unit 824 a for the first transmitantenna 234 a delays the modulated signal by zero seconds, delay unit824 b for the second transmit antenna 234 b delays the modulated signalby one sample period (or T_(sam) seconds), and so on, and delay unit 824t for the T-th transmit antenna 234 t delays the modulated signal by(T−1) sample periods (or (T−1)·T_(sam) seconds). A sample period isequal to T_(sam)=1/(BW·(K+C)), where BW is the overall bandwidth of thesystem in Hertz.

FIG. 8C shows a timing diagram for the T transmissions from the Ttransmit antennas for the embodiments shown in FIGS. 8A and 8B. The sameOFDM symbol is transmitted from each of the T transmit antennas.However, the OFDM symbol sent from each transmit antenna is delayed by adifferent amount. The T delayed and non-delayed OFDM symbols for the Tantennas may be viewed as T different versions of the same OFDM symbol.

For the embodiments shown in equations (7) through (9) and FIGS. 8Athrough 8C, the delays for the T transmit antennas are in integernumbers of sample periods. Phase slopes that result in non-integerdelays for the T transmit antennas (or${{B(i)} = {\mathbb{e}}^{{- {j\pi}}\quad\frac{\mathbb{i}}{L}}},$where L>1) may also be implemented. For example, the time-domain samplesfrom OFDM modulator 230 b in FIG. 8A may be up-sampled to a higher rate(e.g., with a period of T_(upsam)=T_(sam)/L), and the higher ratesamples may be delayed by digital delay units 822 by integer numbers ofthe higher rate sample period (T_(upsam)). Alternatively, analog delayunits 824 in FIG. 8B may provide delays in integer numbers of T_(upsam)(instead of T_(sam)).

When the number of transmit antennas is less than the cyclic prefixlength (or T<C), the cyclic prefix appended to each OFDM symbol makes alinear delay by digital delay units 822 or analog delay units 824appears like a circular rotation for the circular convolution with thetime-domain impulse response g_(i)(n). The weights as defined inequation (7) may thus be implemented by a time delay of i sample periodsfor each transmit antenna i, as shown in FIGS. 8A through 8C. However,as shown in FIG. 8C, the OFDM symbol is transmitted from the T transmitantennas at different delays, which reduces the effectiveness of thecyclic prefix to protect against multipath delay.

The IDFT of K weighted symbols (which are obtained by multiplying Kmodulation symbols with the phase slope shown in equation (7)) providesa time-domain sample sequence that is equal to a circular shift of the Ktime-domain samples from the IDFT of the K (original unweighted)modulation symbols. The spatial processing may thus be performed bycircularly shifting these K time-domain samples.

FIG. 9A shows a block diagram of an OFDM modulator 230 d and a TXspatial processor 220 d, which are yet another embodiment of OFDMmodulator 230 and TX spatial processor 220, respectively, in FIG. 2.Within OFDM modulator 230 d, an IDFT unit 932 performs a K-point IDFT onthe K modulation symbols and provides K time-domain samples, and a P/Sconverter 934 serializes the K time-domain samples. TX spatial processor220 d includes T circular shift units 922 a through 922 t for the Ttransmit antennas. Each unit 922 receives the K time-domain samples fromP/S converter 934, performs a circular shift of the K time-domainsamples by i samples for transmit antenna i, and provides acircular-shifted transformed symbol containing K samples. In particular,unit 922 a performs a circular shift by 0 sample for transmit antenna234 a, unit 922 b performs a circular shift by 1 sample for transmitantenna 234 b, and so on, and unit 922 t performs a circular shift by(T−1) samples for transmit antenna 234 t. T cyclic prefix generators 936a through 936 t receive the circular-shifted transformed symbols fromunits 922 a through 922 t, respectively. Each cyclic prefix generator936 appends a C-sample cyclic prefix to its circular-shifted transformedsymbol and provides an OFDM symbol containing (K+C) samples. Thesubsequent processing by transmitter units 232 a through 232 t is asdescribed above.

FIG. 9B shows a timing diagram for the T transmissions from the Ttransmit antennas for the embodiment shown in FIG. 9A. A differentversion of the OFDM symbol is generated for each of the T transmitantennas by circularly shifting a different amount. However, the Tdifferent versions of the OFDM symbol are sent from the T transmitantennas at the same time.

The embodiments shown in FIGS. 8A, 8B, and 9A illustrate some of theways in which spatial processing for steering diversity may beperformed. In general, the spatial processing for steering diversity maybe performed in various manners and at various locations within thetransmitting entity. For example, the spatial processing may beperformed in the time-domain or the frequency-domain, using digitalcircuitry or analog circuitry, prior to or after the OFDM modulation,and so on.

Equations (6) and (7) represent a function that provides linearlychanging phase shifts across the K subbands for each transmit antenna.The application of linearly changing phase shifts to modulation symbolsin the frequency domain may be achieved by either delaying or circularlyshifting the corresponding time-domain samples, as described above. Ingeneral, the phase shifts across the K subbands for each transmitantenna may be changed in a continuous manner using any function so thatthe beams are varied in a continuous instead of abrupt manner across thesubbands. A linear function of phase shifts is just one example of acontinuous function. The continuous change ensures that the performancefor single-antenna devices that rely on some amounts of correlationacross the subbands (e.g., to simplify channel estimation) is notdegraded.

In the description above, steering diversity is achieved for atransmission of one modulation symbol on each subband in each symbolperiod. Multiple (S) modulation symbols may also be sent via the Ttransmit antennas on one subband in one symbol period to a multi-antennareceiving entity with R receive antennas using steering diversity, whereS≦min{T, R}

The steering diversity techniques described herein may be used forvarious wireless systems. These techniques may also be used for thedownlink (or forward link) as well as the uplink (or reverse link).Steering diversity may be performed by any entity equipped with multipleantennas.

Steering diversity may be used in various manners. For example, atransmitting entity (e.g., an access point or a user terminal) may usesteering diversity to transmit to a receiving entity (e.g., anotheraccess point or user terminal) when accurate information about thewireless channel is not available. Accurate channel information may notbe available due to various reasons such as, for example, a feedbackchannel that is corrupted, a system that is poorly calibrated, thechannel conditions changing too rapidly for the transmitting entity touse/adjust beam steering on time, and so on. The rapidly changingchannel conditions may be due to, for example, the transmitting and/orreceiving entity moving at a high velocity.

Steering diversity may also be used for various applications in awireless system. In one application, broadcast channels in the systemmay be transmitted using steering diversity as described above. The useof steering diversity allows wireless devices in the system to possiblyreceive the broadcast channels with improved reliability, therebyincreasing the range of the broadcast channels. In another application,a paging channel is transmitted using steering diversity. Again,improved reliability and greater coverage may be achieved for the pagingchannel via the use of steering diversity. In yet another application,an 802.11a access point uses steering diversity to improve theperformance of user terminals under its coverage area.

The steering diversity techniques described herein may be implemented byvarious means. For example, these techniques may be implemented inhardware, software, or a combination thereof. For a hardwareimplementation, the processing units used to perform spatial processingfor steering diversity may be implemented within one or more applicationspecific integrated circuits (ASICs), digital signal processors (DSPs),digital signal processing devices (DSPDs), programmable logic devices(PLDs), field programmable gate arrays (FPGAs), processors, controllers,micro-controllers, microprocessors, other electronic units designed toperform the functions described herein, or a combination thereof.

For a software implementation, the steering diversity techniques may beimplemented with modules (e.g., procedures, functions, and so on) thatperform the functions described herein. The software codes may be storedin a memory unit (e.g., memory unit 242 in FIG. 2) and executed by aprocessor (e.g., controller 240). The memory unit may be implementedwithin the processor or external to the processor, in which case it canbe communicatively coupled to the processor via various means as isknown in the art.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. A method of transmitting data in a wireless communication system,comprising: obtaining input symbols to be transmitted on a plurality offrequency subbands of a plurality of antennas; modifying an input symbolfor each frequency subband of each antenna with a phase shift selectedfor the frequency subband and the antenna to generate a phase-shiftedsymbol for the frequency subband and the antenna; and processingphase-shifted symbols for the plurality of frequency subbands of eachantenna to obtain a sequence of samples for the antenna.
 2. The methodof claim 1, further comprising: applying linearly varying phase shiftsacross the plurality of frequency subbands for each antenna.
 3. Themethod of claim 1, further comprising: applying a different phase slopeacross the plurality of frequency subbands for each antenna.
 4. Themethod of claim 1, further comprising: applying continuously varyingphase shifts across the plurality of frequency subbands for eachantenna.
 5. The method of claim 4, further comprising: determining thecontinuously varying phase shifts across the frequency subbands for eachantenna based on a function selected for the antenna.
 6. The method ofclaim 1, wherein the processing the phase-shifted symbols comprisesperforming orthogonal frequency division multiplexing (OFDM) modulationon the phase-shifted symbols for the plurality of frequency subbands ofeach antenna to obtain the sequence of samples for the antenna.
 7. Anapparatus in a wireless communication system, comprising: a spatialprocessor to obtain input symbols to be transmitted on a plurality offrequency subbands of a plurality of antennas and to modify an inputsymbol for each frequency subband of each antenna with a phase shiftselected for the frequency subband and the antenna to generate aphase-shifted symbol for the frequency subband and the antenna; and amodulator to process phase-shifted symbols for the plurality offrequency subbands of each antenna to obtain a sequence of samples forthe antenna.
 8. The apparatus of claim 7, wherein the spatial processorapplies linearly varying phase shifts across the plurality of frequencysubbands for each antenna.
 9. The apparatus of claim 7, wherein thespatial processor applies a different phase slope across the pluralityof frequency subbands for each antenna.
 10. The apparatus of claim 7,wherein the spatial processor applies continuously varying phase shiftsacross the plurality of frequency subbands for each antenna.
 11. Anapparatus in a wireless communication system, comprising: means forobtaining input symbols to be transmitted on a plurality of frequencysubbands of a plurality of antennas; means for modifying an input symbolfor each frequency subband of each antenna with a phase shift selectedfor the frequency subband and the antenna to generate a phase-shiftedsymbol for the frequency subband and the antenna; and means forprocessing phase-shifted symbols for the plurality of frequency subbandsof each antenna to obtain a sequence of samples for the antenna.
 12. Theapparatus of claim 11, further comprising: means for applying linearlyvarying phase shifts across the plurality of frequency subbands for eachantenna.
 13. The apparatus of claim 11, further comprising: means forapplying a different phase slope across the plurality of frequencysubbands for each antenna.
 14. The apparatus of claim 11, furthercomprising: means for applying continuously varying phase shifts acrossthe plurality of frequency subbands for each antenna.
 15. A method oftransmitting data in a wireless communication system, comprising:processing data to obtain an input sequence of time-domain samples;generating a plurality of output sequences of time-domain samples for aplurality of antennas by temporally modifying the input sequence oftime-domain samples; and transmitting the plurality of output sequencesfrom the plurality of antennas.
 16. The method of claim 15, wherein theprocessing the data comprises performing an inverse discrete Fouriertransform on a plurality of input symbols for a plurality of frequencysubbands to obtain a plurality of time-domain samples, and repeating aportion of the plurality of time-domain samples to obtain the inputsequence of time-domain samples.
 17. The method of claim 16, wherein thegenerating the plurality of output sequences of time-domain samplescomprises delaying the input sequence by different amounts to generatethe plurality of output sequences.
 18. The method of claim 16, whereinthe generating the plurality of output sequences of time-domain samplescomprises delaying the input sequence by different integer numbers ofsample periods to generate the plurality of output sequences.
 19. Themethod of claim 16, wherein the generating the plurality of outputsequences of time-domain samples comprises delaying the input sequenceby different fractional amounts of a sample period to generate theplurality of output sequences.
 20. The method of claim 16, wherein thetransmitting the plurality of output sequences comprises transmittingthe plurality of output sequences from the plurality of antennasstarting at different times.
 21. The method of claim 15, wherein theprocessing the data comprises performing an inverse discrete Fouriertransform on a plurality of input symbols for a plurality of frequencysubbands to obtain the input sequence of time-domain samples.
 22. Themethod of claim 21, wherein the generating the plurality of outputsequences of time-domain samples comprises circularly shifting the inputsequence of time-domain samples by different amounts to obtain aplurality of intermediate sequences of time-domain samples, andrepeating a portion of each intermediate sequence of time-domain samplesto obtain a respective output sequence of time-domain samples.
 23. Themethod of claim 22, wherein the circularly shifting the input sequencecomprises circularly shifting the input sequence by different integernumbers of samples to obtain the plurality of intermediate sequences.24. The method of claim 21, wherein the transmitting the plurality ofoutput sequences comprises transmitting the plurality of outputsequences from the plurality of antennas starting at same time.
 25. Anapparatus in a wireless communication system, comprising: a modulator toprocess data to obtain an input sequence of time-domain samples; aprocessor to generate a plurality of output sequences of time-domainsamples for a plurality of antennas by temporally modifying the inputsequence of time-domain samples; and a plurality of transmitter units totransmit the plurality of output sequences from the plurality ofantennas.
 26. The apparatus of claim 25, wherein the modulator performsan inverse discrete Fourier transform on a plurality of input symbolsfor a plurality of frequency subbands to obtain a plurality oftime-domain samples and further repeats a portion of the plurality oftime-domain samples to obtain the input sequence of time-domain samples.27. The apparatus of claim 26, wherein the processor delays the inputsequence by different amounts to generate the plurality of outputsequences.
 28. The apparatus of claim 26, wherein the processorcomprises a plurality of delay unit to delay the input sequence bydifferent fractional amounts of a sample period to generate theplurality of output sequences.
 29. The apparatus of claim 25, whereinthe plurality of transmitter units transmit the plurality of outputsequences from the plurality of antennas starting at different times.30. The apparatus of claim 25, wherein the processor circularly shiftsthe input sequence of time-domain samples by different amounts to obtaina plurality of intermediate sequences of time-domain samples and repeatsa portion of each intermediate sequence of time-domain samples to obtaina respective output sequence of time-domain samples.
 31. The apparatusof claim 25, wherein the plurality of transmitter units transmit theplurality of output sequences from the plurality of antennas starting atsame time.
 32. An apparatus in a wireless communication system,comprising: means for processing data to obtain an input sequence oftime-domain samples; means for generating a plurality of outputsequences of time-domain samples for a plurality of antennas bytemporally modifying the input sequence of time-domain samples; andmeans for transmitting the plurality of output sequences from theplurality of antennas.
 33. The apparatus of claim 32, wherein the meansfor processing the data comprises means for performing an inversediscrete Fourier transform on a plurality of input symbols for aplurality of frequency subbands to obtain a plurality of time-domainsamples, and means for repeating a portion of the plurality oftime-domain samples to obtain the input sequence of time-domain samples.34. The apparatus of claim 33, wherein the means for generating theplurality of output sequences of time-domain samples comprises means fordelaying the input sequence by different amounts to generate theplurality of output sequences.
 35. The apparatus of claim 32, whereinthe means for transmitting the plurality of output sequences comprisesmeans for transmitting the plurality of output sequences from theplurality of antennas starting at different times.
 36. The apparatus ofclaim 32, wherein the means for generating the plurality of outputsequences of time-domain samples comprises means for circularly shiftingthe input sequence of time-domain samples by different amounts to obtaina plurality of intermediate sequences of time-domain samples, and meansfor repeating a portion of each intermediate sequence of time-domainsamples to obtain a respective output sequence of time-domain samples.37. The apparatus of claim 32, wherein the means for transmitting theplurality of output sequences comprises means for transmitting theplurality of output sequences from the plurality of antennas starting atsame time.